AN75
AN75
High Power Factor LED Replacement T8 Fluorescent Tube
using the AL9910 High Voltage LED Controller
Yong Ang, Diodes Inc.
Introduction
This application note describes the principles and design equations required for the design of a high
brightness LED lamp using the AL9910. The equations are then used to demonstrate the design of a
universal, offline, high power factor (PF), 13W LED lamp suitable for use as the replacement for T8
fluorescent tube. A complete design including the electrical diagram, component list and performance
measurements are provided.
AL9910 high power factor buck LED driver
Figure 1 Electrical schematic of a high power factor 13W LED lamp
Figure 1 shows the electrical diagram of an offline 13W LED driver.
On the input side, CX1, CX2, CX3, CX4, L1 and L2 provide sufficient filtering for both differential mode
and common mode EMI noise which are generated by the switching converter circuit.
The rectified AC line voltage from the bridge rectifier DB1 is then fed into a passive power factor
correction or valley fill circuit which consists of 3 diodes and 2 capacitors. D1, D2, D3, C1, C2 improve
the input line current distortion in order to achieve PF greater than 0.9 for the AC line input.
The constant current regulator section consists of a buck converter driven by the AL9910. Normally,
the buck regulator is used in fixed frequency mode but its duty cycle limitation of 50% is not practical
for offline lamp. This problem can be overcome by changing the control method to a fixed off-time
operation.
The design of the internal oscillator in the AL9910 allows the IC to be configured for either fixed
frequency or fixed off-time based on how resistor R
T
is connected. For fixed off-time operation, the
resistor R
T
is connected between the Gate and R
OSC
pins, as shown in Figure 1. This converter has
now a constant off-time when the power MOSFET is turned off. The on-time is based on the current
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sense signal and the switching adjusts to be the sum of the on- and off-time. This change allows the
converter to work with duty cycles greater than 50%.
Design Guide – High power factor offline LED driver
In this section the design procedure is outlined according to the schematic shown in Figure 1. First,
the guideline for selecting the components for valley fill power factor correction stage and fixed off-
time buck converter is shown. The power inductor calculation is then demonstrated and finally, the
power losses within MOSFET and free-wheel diode are assessed.
The specifications for the system are:
V
AC
= 230Vac
V
AC(min)
= 85Vac
V
AC(max)
= 264Vac
I
LED(nom)
= 240mA
V
LED(nom)
= 54V
V
LED(min)
= 42V
V
LED(max)
= 59V
P
OUT
= 12.96W
f
swi(nom)
= 55kHz
Passive factor correction stage design
The purpose of the valley fill circuit (see Figure 2) is to allow the buck converter to pull power directly
off the AC line when the line voltage is greater than 50% of its peak voltage.
Figure 2 Valley-fill PFC stage and operating waveforms (Green: V
IN
to LED driver; Orange:
AL9910’s gate voltage)
The maximum bus voltage at the input of the buck converter is,
V
IN(max)
=
2
×
V
ac(max)
=
2
×
264 Vac
=
373 V
During this time, capacitors within the valley fill circuit (C1 and C2) are in series and charged via D2
and R1. If the capacitors have identical capacitance value, the peak voltage across C1 and C2
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is
V
IN(max)
2
=
186 V
. Often a 20% difference in capacitance could be observed between like
capacitors. Therefore a voltage rating margin of 25% should be considered.
Once the line drops below 50% of its peak voltage, the two capacitors are essentially placed in
parallel. The bus voltage V
IN(min)
is the lowest voltage value at the input of the buck converter. V
IN(min)
at the minimum AC line voltage V
ac(min)
is,
V
IN(min)
=
2
×
V
ac(min)
2
=
2
×
85 Vac 2
=
60 V
At 60Hz, the total time of a half AC line cycle is 8.33ms. The power to the buck converter is derived
from the valley-fill capacitors when the AC line voltage is equal to or less than 50% of its peak voltage.
The hold up time for the capacitors equates to
t
HOLD
=
1 3
×
8.33ms
=
2.77ms
. The valley-fill capacitor
value can then be calculated,
P
out
C
TOTAL
=
V
IN(min)
×
t
HOLD
=
12.96 W
×
2.77ms
60 V
=
30
μ
F
20 V
V
DROOP
Therefore,
C1
=
C2
=
15
μ
F
. V
DROOP
is the voltage droop on the capacitors when they are delivering full
power to the buck converter. Ideally V
DROOP
should be set to less than
V
DROOP
=
V
IN(min)
−
V
LED(max)
in
order to ensure continuous LED conduction at low line voltage. Nevertheless, V
DROOP
is set to be 20V
in the design example to avoid the need for very large valley-fill electrolytic capacitor.
A 20V V
DROOP
implies that the bus voltage V
IN
at the input of buck converter will drop to 40V during
part of the AC line cycle. As the buck regulator requires V
IN
to be greater than the LED stack voltage
(V
LED(max)
=59V) for regulation, the LED will be off during part of the AC line cycle. This has the effect of
reducing the actual output LED current at low AC input voltage. In the design example, the LED
current drops by approximately 20% from its nominal value at 85Vac (see Figure 4).
Setting the fixed off-time and switching frequency range
For fixed off-time operation, the switching frequency will vary subjected to the actual input voltage and
output LED conditions.
A nominal switching frequency f
swi(nom)
should be chosen. A high nominal switching frequency will
result in smaller inductor size, but could lead to increased switching losses in the circuit. A good
design practice is to choose a nominal switching frequency knowing that the switching frequency will
decrease as the line voltage drops and increases as the line voltage increases.
The fixed off-time t
OFF
can be computed as,
1-
t
off
=
V
LED(nom)
V
ac(nom)
f
swi(nom)
54V
=
230V
=
13.9
μ
s
55kHz
1-
The off-time is programmed by timing resistor R
T
as shown in Figure 1. The value of R
T
is given by,
R
T
(
k
Ω
)
=
t
OFF
(
μ
s
)
×
25
−
22
=
13.9
×
25
−
22
=
326k
Ω
A 330kΩ is selected for R
T
. Next, the two extremes of the variable switching frequency can be
approximated as,
f
swi(min)
=
f
swi(max)
=
1
−
V
LED(max)
V
IN(min)
t
OFF
1
−
V
LED(min)
V
IN(max)
t
OFF
=
=
1
−
59 V 69 V
=
10kHz
13.9
μ
s
1
−
42V 373 V
=
63.8kHz
13.9
μ
s
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It is advisable to keep below the maximum switching frequency f
swi(max)
below 150kHz to avoid
excessive switching loss.
Inductor selection and setting the LED current
The fixed off-time architecture of the AL9910 regulates the average current through the inductor L
BUCK
.
The value of L
BUCK
depends on the desirable peak-to-peak ripple
ΔI
L
in the output LED current. L
BUCK
can be set with the following equation,
L
BUCK
=
V
LED(nom )
×
t
OFF
Δ
I
L
=
54 V
×
13.9
μ
s
=
6.6mH
115mA
Due to diameter limitation of the T8 tube, L
BUCK
is made up of L3 and L4 as shown in Figure 1.
The AL9910 constant off-time control loop regulates the peak inductor current I
pk
. As the average
inductor current equals the average LED current, the average LED current can be regulated by
controlling I
pk
.
Given a fixed inductor value, the change in the inductor current over time is proportional to the voltage
applied across the inductor. During the off-time, the voltage seen by the inductor is the LED stack
voltage. So, the peak inductor current should be regulated to,
I
pk
=
I
LED(nom)
+
0.5
×
V
LED(nom )
×
t
OFF
L
BUCK
=
240mA
+
0.5
×
54 V
×
13.9
μ
s
=
297mA
6.6mH
The peak current is constant and set by the sense resistor R
SENSE
. If the LD pin is tied to the VDD pin,
the value of R
SENSE
can be easily calculated because the voltage threshold on the CS pin is 0.25V,
R
SENSE
=
0.25
=
0.84
Ω
297mA
In the circuit shown in Figure 1, R
SENSE
consists of R5, R6 and R7.
The peak current rating of the L
BUCK
should be greater than I
pk
and the RMS current rating of the
inductor should be at least 110% of I
LED(nom)
.
Although the described solution, working in fixed off-time and Continuous Conduction Mode (CCM),
works as a constant current source, a limitation to the output LED current accuracy is its dependency
on the number of LEDs and overall LED chain voltage. The best result can be achieved using a fixed
number of LEDs. A variable number of LEDs results in reduced current precision.
The two extremes of the output LED current can be approximated as,
I
LED(min)
=
I
pk
-
0.5
×
V
LED(max)
×
t
OFF
L
BUCK
=
297mA -
0.5
×
59 V
×
13.9
μ
s
=
234mA
6.6mH
I
LED(max)
=
I
pk
-
0.5
×
V
LED(min)
×
t
OFF
L
BUCK
=
297mA -
0.5
×
42V
×
13.9
μ
s
=
253mA
6.6mH
The above equation shows that the precision of the LED current also depends on the tolerance of
practical inductor L
BUCK
. Inductor with tolerance rating equal or less than 10% should be chosen to
ensure good LED current precision at mass production.
Power MOSFET calculation
The power MOSFET is chosen based on maximum voltage stress, peak MOSFET current, total power
losses, maximum allowable working temperature and the gate driver capability of the AL9910.
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Maximum drain-source voltage stress on the power MOSFET for this converter is equal to the input
voltage. However, a typical voltage safety margin for the MOSFET defines the maximum reverse
voltage as follows,
V
DSS
=
1.3
×
V
IN(max)
=
1.3
×
373 V
=
485 V
which implies that a common 500V MOSFET is suitable.
The power MOSFET losses will be dominated by switching loss. The switching loss depends on the
switching time, frequency, MOSFET drain current and drain-source voltage. The switching rise time
t
RISE
and fall time t
FALL
is a function of the MOSFET’s gate capacitance, the gate driver capability of the
AL9910 and layout design. The worse case switching power losses occurs at V
LED(min)
and V
IN(max)
.
The switching loss is approximately,
V
t
⎛
⎞
V
IN(max)
× ⎜
I
pk
−
LED(min) OFF
⎟ ×
t
RISE
×
f
swi(max)
⎜
⎟
V
IN(max)
×
I
pk
×
t
FALL
×
f
swi(max)
L
BUCK
⎝
⎠
=
+
2
2
373V
×
(
297mA
−
88mA
)
×
65ns
×
63.8kHz 373V
×
65ns
×
63.8kHz
=
+
2
2
=
455mW
P
SW
where the switching time t
RISE
and t
FALL
are measured to be 65ns with the 600V MOSFET
SPB03N60S5 as the power MOSFET. As shown in Figure 1, R10 is a series gate resistor that slows
down the MOSFET switching and reduces EMI emission.
The RMS current through the MOSFET at V
LED(min)
and V
IN(max)
is given by,
I
D(RMS)
=
=
V
LED(min)
V
IN(max)
V
LED(min)
×
t
OFF
L
BUCK
⎛
× ⎜
I
LED(nom )
+
⎜
12
⎝
⎞
⎟
⎟
⎠
42V
⎛
42V
×
13.9
μ
s 6.6mH
⎞
⎟
× ⎜
240mA
+
⎟
373 V
⎜
12
⎝
⎠
=
89mA
The power MOSFET conduction loss depends on its static drain-source resistance R
DS(ON)
at the
MOSFET working temperature. It is possible to calculate the continuous conduction loss:
2
P
COND
=
I
D(RMS)
×
R
DS(ON)
=
(
89mA
)
×
2.5
Ω =
19mW
2
The total power MOSFET loss is:
P
TOT
=
P
SW
+
P
COND
=
455mW
+
19mW
=
474mW
Total MOSFET power loss is dissipated from the SMD package into the PC Board. So it is possible to
calculate the MOSFET working junction temperature can be calculated if the package junction-to-
ambient thermal resistance R
thJA
is known. The calculated MOSFET junction temperature, T
J
, must be
lower then the maximum allowable junction temperature T
J(MAX)
:
T
J
=
P
TOT
×
θ
thJA
+
T
AMB
=
474mW
×
62
o
C W
+
80
o
C
=
109.4
o
C
The internal ambient temperature within the LED converter, T
AMB
, is assumed to be 80ºC.
θ
thJA
=
62
o
C W
is the thermal resistance for TO-263 with minimum copper area. For practical design, it is
recommended to keep the junction temperature below 110ºC to avoid temperature stress on the
device.
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