Data Sheet No. PD94703
IRU3065
POSITIVE TO NEGATIVE DC TO DC CONTROLLER
PRODUCT DATASHEET
FEATURES
Generate Negative Output from +5V Input
1A Maximum Output Current
1.5MHz maximum Switching Frequency
Few External Components
Available in 6-Pin SOT-23
DESCRIPTION
The IRU3065 controller is designed to provide solutions
for the applications requiring low power on board switch-
ing regulators. The IRU3065 is specifically designed
for positive to negative conversion and uses few com-
ponents for a simple solution. The IRU3065 operates at
high switching frequency (up to 1.5MHz), resulting in
smaller magnetics. The output voltage can be set by
using an external resistor divider. The stability over all
conditions is inherent with this architecture without any
compensation. The device is available in the standard
6-Pin SOT-23.
APPLICATIONS
Hard Disk Drives
Blue Laser for DVD R-W
MR Head Bias
LCD Bias
GaAs FET Bias
Positive-to-Negative Conversion
TYPICAL APPLICATION
5V
D1
BAT54
V
DD
Vcc
C3
100pF
C1 1uF
C4
10uF
U1
IRU3065
V
GATE
Gnd
Q1
IRLML5203
D2
C6
10uF
V
OUT
(-5V)
10BQ015
L1
1.2uH
R1
0.1
R3
10K
V
SEN
R2
10K
I
SEN
V
REF
= 5V
V
OUT
= -V
REF
3
R3
R2
Figure 1 - Typical application of IRU3065 for single input voltage.
PACKAGE ORDER INFORMATION
T
A
(°C)
0 To 70
DEVICE
IRU3065CL
PACKAGE
6-Pin SOT-23 (L6)
OUTPUT VOLTAGE
Adjustable
Rev. 1.0
11/10/04
www.irf.com
1
IRU3065
ABSOLUTE MAXIMUM RATINGS
Vcc ......................................................................... 7V
V
DD
......................................................................... 12V
Operating Junction Temperature Range ..................... 0°C To 125°C
Operating Ambient Temperature Range ..................... 0°C To 70°C
Storage Temperature Range ...................................... -65°C To +150°C
ESD Capability (Human Body Model) ........................ 2000V
PACKAGE INFORMATION
6-PIN PLASTIC SOT-23 (L6)
TOP VIEW
V
GATE
1
Gnd 2
V
SEN
3
6 Vcc
5 V
DD
4 I
SEN
υ
JA
=2308C/W
ELECTRICAL SPECIFICATIONS
Unless otherwise specified, these specifications apply over Vcc=5V, V
DD
=7V, C
GATE
=470pF, R
SEN
=0.125ς,
R
FDBK1
=R
FDBK2
=10Kς (to Vcc), fs=1.2MHz, I
FL
=0.25A and T
J
=0°C to 125°C. Typical values refer to T
J
=25°C.
PARAMETER
SYM
TEST CONDITION
Recommended Vcc Supply
Vcc Note.1
Recommended V
DD
Supply
V
DD
Operating Current
Icc
Initial Output Voltage Accuracy
Measured in application
T
J
=258C, Vout=-5V
Output Accuracy
Measured in application
over temp. Vout=-5V.
Voltage Feedback Sense
V
V
SEN
Voltage Feedback Input Offset V
V
off
Voltage Feedback Bias Current I
V
BIAS
Peak Current Sense Voltage
V
I
s
Min Current Sense Voltage
V
I
s
Current Sense Bias Current
I
I
BIAS
Output Drivers Section
Switching Frequency
Note. 1
fs
Max Output Duty Cycle
Dmax
Min Output Duty Cycle
Dmin
10% to 90% Vgate high
Rise Time
Tr
Fall Time
T
f
90% to 10% Vgate going low
Propagation Delay from
T
D
Vsens=1V. Isens from 0 to
250mV. Delay time between
Current Sense to Output
90% of Isens to 10% of Vgate
MIN
4
4
-1%
-2%
0
-10
145
50
2
1.5
100
0
40
40
100
10
2
TYP
5
3
1%
+2%
V
mV
µA
mV
mV
µA
MHz
%
%
ns
ns
ns
MAX
UNITS
V
V
mA
Note. 1. guarantted by design
2
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Rev. 1.0
11/10/04
IRU3065
PIN DESCRIPTIONS
PIN#
1
2
3
4
5
6
PIN SYMBOL
V
GATE
Gnd
V
SEN
I
SEN
V
DD
Vcc
PIN DESCRIPTION
Output driver for external P Channel MOSFET.
This pin serves as ground pin and must be connected to the ground plane.
A resistor divider from this pin to V
OUT
and Vcc or an external V
REF
, sets the output
voltage.
This pin sets the maximum load current by sensing the inductor current.
This pin provides biasing for the output driver.
This pin provides biasing for the internal blocks of the IC.
BLOCK DIAGRAM
Vcc
6
V
DD
5
S
Q
R
1 V
GATE
4 I
SEN
3 V
SEN
2
Gnd
Figure 2 - Simplified block diagram of the IRU3065.
Rev. 1.0
11/10/04
www.irf.com
3
IRU3065
APPLICATION INFORMATION
Introduction
The IRU3065 is a controller intended for an inverting
regulator solution. For example, to generate –5V from
a 5V supply. The controller is simple and only has a
voltage comparator, current hysteretic comparator, flip-
flop and MOSFET driver. It controls a typical buck boost
converter configured by a P-channel MOSFET, an in-
ductor, a diode and an output capacitor. The sensed
inductor current by a sensing resistor compares with
current comparator. The current comparator uses hys-
teresis to control the turn-on and turn-off of the transis-
tor based upon the inductor current and gated by the
output voltage level. When the inductor current rises
past the hysteresis set point, the output of the current
comparator goes high. The flip-flop is reset and the P-
channel MOSFET is turned off. In the mean time, the
current sense reference is reduced to near zero, giving
a zero reference threshold voltage level. As the induc-
tor current passes below this threshold, which indicates
that the inductor’s stored energy has been transferred
to the output capacitor, the current comparator output
goes high and turns on the output transistor (if the out-
put voltage is low). By means of hysteresis, the induc-
tor charges and discharges and functions as self oscil-
lating. The voltage feedback comparator acts as a de-
mand governor to maintain the output voltage at the de-
sired level.
By hysteresis control, the maximum switch current (also
equals inductor current) is limited by the internal cur-
rent sensing reference. The power limit is automatically
achieved. The switching frequency is determined by a
combination of factors including the inductance, output
load current level and peak inductor current. The theo-
retical output voltage and switching frequency versus
output current is shown in Figure 3.
Output
voltage
Regulation
mode
Power limit
mode
When the output current is below a critical current I
OCP
,
the output voltage is regulated at the desired value and
the switching frequency increases as output current
increases. At current I
OCP
, the switching frequency
reaches its maximum f
S(MAX)
. In this region, the opera-
tion is in regulation mode. When the current goes above
I
OCP
, the operation goes into power limit mode. The out-
put voltage starts to decrease and the output power is
limited. The switching frequency is also reduced.
Analysis shows that the current I
OCP
is determined by:
VI
SEN(TH)
V
IN
I
OCP
= 1
3
3
Rs
V
IN
-V
OUT(NOM)
+V
D
2
--(1)
Where:
Rs = Current Sensing Resistance
VI
SEN(TH)
= Upper Threshold Voltage at the current
comparator (when Vcc=5V, VI
SEN(TH)
=0.145V)
V
IN
= Input Voltage
V
D
= Diode Forward Voltage
V
OUT(NOM)
= Nominal Output Voltage
The maximum switching frequency is determined by:
f
S(MAX)
=
f
S(MAX)
=
V
IN
3(V
D
-V
OUT(NOM)
)
(V
IN
+V
D
-V
OUT(NOM
)3L3I
PEAK
V
IN
3(V
D
-V
OUT(NOM)
)3R
S
VI
SEN(TH)
3(V
IN
+V
D
-V
OUT(NOM)
)3L
---(2)
Where:
I
PEAK
= Peak Inductor Current
I
PEAK
is determined by:
I
PEAK
=
VI
SEN(TH)
R
S
---(3)
The detailed operation can be seen in the theoretical
operation section
V
out
f
s max
Switching
frequency
I
out
f
s
I
out
I
ocp
Figure 3 - Theoretical output voltage and switching
frequency vs. output current.
4
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Rev. 1.0
11/10/04
IRU3065
APPLICATION EXAMPLE
Design Example
The following design example is for the evaluation board
application for IRU3065. The schematic is shown in fig-
ure 1:
Where:
V
IN
= 5V
V
OUT(NOM)
= -5V
I
OUT
= 200mA
f
S(MAX)
= Maximum Frequency
f
S(MAX)
= 1.2MHz
V
D
= Diode Forward Voltage
V
D
= 0.5V
Vcc = 5V
VI
SEN(TH)
=145mV
≅
150mV
Voltage Sensing Resistor
The output voltage is determined by the two voltage sens-
ing resistors R2 and R3:
V
OUT(NOM)
= - R3
3
V
REF
R2
If R3 is chosen as 10K, Then R2 is given by:
V
REF
5V
R2 = -
3
R3 = -
3
10K = 10Kς
V
OUT(NOM)
-5V
Current Sensing Resistor R
S
In order to select R
S
, the desired critical current I
OCP
has to be determined. Considering the switching losses,
for conservative, the critical current should select to be
slightly greater than the nominal output current.
Select:
I
OCP
= 200mA31.5 = 300mA
Where 1.5 is the coefficient to take the efficiency into
account.
According to equation (1), the current I
OCP
is given by:
I
OCP
=
V
IN
1 0.15
3
3
= 300mA
R
S
V
IN
- V
OUT(NOM)
+ V
D
2
V
IN
1 0.15
3
3
2 I
OCP
V
IN
- V
OUT(NOM)
+ V
D
ψ
0.12ς
The modified current I
OCP
is:
I
OCP
=
I
OCP
0.15
V
IN
1
3
3
R
S
V
IN
+ V
D
- V
OUT(NOM)
2
5
1
=
3
3
1.5A = 357mA
5 + 0.5 - (-5)
2
Output Inductor L
The inductance is chosen by equation (2):
L
]
V
IN
3(V
D
- V
OUT(NOM)
)
(V
IN
+V
D
-
V
OUT(NOM)
)3f
S(MAX)
3I
PEAK
-(-5 - 0.5)
5
L
]
3
1.5A (5 - (-5) + 0.5)31.2MHz
= 1.45µH
Select L = 1.2µH
The maximum inductor current is: I
PEAK
= 1.5A
The maximum average inductor current equals
I
AVG
=(VI
SENTH_MAX
+VI
SENTH_MIN
)/Rs/2
I
AVG
=(145mV+50mV)/0.1ohm/2=1A
MOSFET Selection
A P-channel MOSFET is required. The peak current in
this case is equal to I
PEAK
=1.5A. The MOSFET
IRLML5203, from international Rectifier with I
D
=3A and
BVDSS=30V, is a good choice.
Input Capacitor
An input capacitor will help to minimize the induced
ripple on the +5V supply. A 1µF to 10µF X7R ceramic
capacitor is recommended.
Output Capacitor
An output capacitor is required to store energy from
transfer to the output inductor. Its capacitance and ESR
have a great impact on output voltage ripple. A 10µF to
22µF X7R Tantolum or ceramic capacitor is recom-
mended.
Output Diode
The average diode current equals output current. In
this case, select the diode average current larger than
300mA. The lowest block voltage is V
IN
+(-V
OUT
). In this
case, It is 10V. In order to reduce the switching losses,
the Schottky diode is recommended. The diode 10BQ015
from International Rectifier with I
D
=1A and V
BR
=15V is
a good choice.
Other Components
In order to speed up the turn off of P-channel MOSFET,
a fast diode 1N4148 or a 100ohm resistor and 100pF
capacitor is connected to the pin V
DD
and V
GATE
as shown
The current sensing resistance is calculated as:
R
S
=
R
S
=
5
1 0.15
3
3
0.3 5 - (-5) + 0.5
2
Select R
S
= 0.1Ω
From equation (3), the modified inductor peak current
is:
VI
SEN(TH)
I
PEAK
=
= 1.5A
R
S
Rev. 1.0
11/10/04
www.irf.com
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