AN1262
APPLICATION NOTE
OFFLINE FLYBACK CONVERTERS DESIGN
METHODOLOGY WITH THE L6590 FAMILY
by Claudio Adragna
The design of flyback converters is quite a demanding task that requires SMPS engineers to cope with sev-
eral problem areas such as magnetics, control loop analysis, power devices, as well as regulations concern-
ing safety, EMC and the emerging standby consumption requirements. Lots of variable are involved and
complex tradeoffs are necessary to meet the goal.
In this scenario, the high-voltage monolithic switchers of the L6590 family greatly simplify the task and, at
the same time, allow to build robust and cost-effective low-power systems.
In this application note, after a review of flyback topology, a step-by-step design procedure of an offline sin-
gle-output flyback converter will be outlined. As an example, the design of the test board will be carried out
in details.
1
FLYBACK BASICS
Flyback operation will be illustrated with reference to the basic circuit and the waveforms of fig. 1. It is a two-
step process. During the ON-time of the switch, energy is taken from the input and stored in the primary winding
of the flyback transformer (actually, two coupled inductors). At the secondary side, the catch diode is reverse-
biased, thus the load is being supplied by the energy stored in the output bulk capacitor.
Figure 1. Flyback Topology and associated waveforms.
Vin
Is
Lp
Vout
Vac
n:1
Vcc
DRAIN
Ls
L6590
L6590D
L6590A
OSCILLATOR
2.5 V
Clock
Ip
Max. Duty cycle
S
R
Driver
Q
1
1/100
ISOLATED
FEEDBACK
+
VFB
-
E/A
+
PWM
-
Clock
LEB
+
OCP
-
Rsense
0.5 V
COMP
GND
FREQUENCY
COMPENSATION
CLOCK
CLOCK
CLOCK
Q
Q
Q
Ip
Ip
Ip
∆Ip
Is
Is
Is
Vdrain
Vin
n•Vout
Vdrain
Vdrain
DCM operation
TRANSITION
CCM operation
May 2001
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AN1262 APPLICATION NOTE
When the switch turns off, the primary circuit is open and the energy stored in the primary is transferred to the
secondary by magnetic coupling. The catch diode is forward-biased, and the stored energy is delivered to the
output capacitor and the load. The output voltage V
out
is reflected back to the primary through the turns ratio n
(V
R
, reflected voltage) and adds up to the input voltage V
in
, giving origin to a much higher voltage on the drain
of the MOSFET.
Flyback is operated in DCM (Discontinuous Conduction Mode) when the input -or primary - current starts from
zero at the beginning of each switching cycle. This happens because the secondary of the transformer has dis-
charged all the energy stored in the previous period. If this energy transfer is not complete, the primary current
will start from a value greater than zero at the beginning of each cycle. Then flyback is said to be operated in
CCM (Continuous Conduction Mode). DCM is characterized by currents shaped in a triangular fashion, whereas
CCM features trapezoidal currents.
The boundary between these two types of operation depends on several parameters. For a given converter,
that is, as the switching frequency, inductance of the primary winding, transformer turns ratio and regulated out-
put voltage are defined, it depends on the input voltage and the output load.
At design time, whether the converter will be operated in CCM or in DCM and where the boundary will be located is
up to the designer. Usually CCM is selected with the objective of maximizing converter's power capability or minimiz-
ing primary RMS current. However, in CCM operation the system's dynamic behavior is considerably worse.
Usually, the converters based on the L6590 family devices are able to deliver the desired output power even
with DCM operation, thus CCM will not be considered.
Table 1. Converter specification data and pre-design choices
Converter Electrical Specification
V
ACmin
V
ACmax
f
L
N
H
V
out
∆V
out%
V
r%
P
outmax
η
T
amb
Minimum mains voltage
Maximum mains voltage
Mains frequency (@ min. mains)
Number of holdup cycles
Regulated output voltage
Percent output voltage tolerance
(±)
Percent output voltage ripple
Maximum output power
Expected converter efficiency
Maximum ambient temperature
Pre-design Choices
V
R
η
T
V
spike
V
cc
V
F
V
BF
2/42
Reflected voltage
Transformer efficiency
Leakage inductance overvoltage
IC supply voltage
Secondary diode forward drop
Bridge Rectifier + EMI filter voltage drop
AN1262 APPLICATION NOTE
2
CONVERTER ELECTRICAL SPECIFICATION
The starting point of the design procedure is the properties of the converter as a black-box, that is the set of data
listed in the electrical specification table (table 1). Additional requirements, such as efficiency at zero load or
line/load regulation or maximum junction temperature, etc., can be added to that list and their impact will be con-
sidered where appropriate.
s
Mains Voltage: Range and Frequency.
There are basically the three possible options listed in table 2,
where a variation of ± 20% is assumed, according to common practice. There are exceptions like some
distribution lines rated at 277 V
AC
, where a ± 10% spread can be considered, or other special cases for
specific applications. Table 2 shows also the line frequency to be considered in the standard cases at
the minimum specified mains voltages. An additional specification may require the converter to be shut
down if the mains voltage falls below a "brownout level". This additional specification will be used for
setting up the brownout protection on the types where it is available.
Table 2. Mains voltage specifications
Input (V
AC
)
110
220
WRM (Wide Range Mains)
s
V
ACmin
(V
AC
)
88
176
88
V
ACmax
(V
AC
)
132
264
264
f
L
(Hz)
60
50
60
Number of holdup cycles.
The holdup requirement is the ability of the converter to keep the output volt-
age in regulation even in case of mains interruption (missing cycles). This is usually specified in terms
of number of mains cycles N
H
. This feature is not always demanded (in which case, N
H
= 0), otherwise
the typical requirement is 1 mains cycle, that is N
H
= 1. It impacts on the input bulk capacitor selection.
Output voltage tolerance.
It can be expressed either in absolute value or as a percentage of the nominal
voltage. This requirement, as well as the ones on line and load regulation, if specified, will affect the
choice of the feedback technique (primary or secondary).
Output voltage ripple.
The ripple superimposed on top of the DC output voltage is specified as the peak-
to-peak amplitude and includes both low frequency (at 2·f
L
) and high frequency (f
sw
) component.
Switching noise due to parasitics of the printed circuit board and random noise are beyond the scope
of this procedure. This requirement, if tight, may require the use of an additional filtering cell at the out-
put.
Converter Efficiency.
The efficiency is, by definition, the ratio of the output power to the input power.
This figure is strongly dependent on the output voltage, because of the losses on the secondary diode.
It should be set based on experience, using numbers of similar converters as a reference. As a rule of
thumb, 75% (η = 0.75) can be used for a low voltage output (3.3 V or 5 V) and 80% (η = 0.8) for higher
output voltages (12 V and above).
PRE-DESIGN CHOICES
s
s
s
3
Before starting the design calculations of the various parts of the converter, some parameters not defined at the
"black-box level" need to be fixed. There is some degree of freedom in the selection of these parameters, pro-
vided some constraints are taken into account.
s
Reflected Voltage.
In principle, the reflected voltage should be as high as possible. In fact this leads to
a greater duty cycle, which minimizes the RMS current through the IC's MOSFET for a given power
throughput. There are two possible limitations to the maximum reflected voltage. One is the maximum
duty cycle D
max
allowed by the devices (67% min.); some margin should be considered for load tran-
sients, thus the reflected voltage should be such that the maximum duty cycle (at minimum input voltage
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AN1262 APPLICATION NOTE
and maximum output power) does not exceed 62-64%. The other limitation is that the sum of the max-
imum input voltage, reflected voltage and overvoltage spike - due to the leakage inductance - must be
below the breakdown of the internal MOSFET (700 V min.). Some margin needs also to be considered:
at least 50V is recommended to take the forward recovery of the diode of the clamp circuit and param-
eter spread into account. Figure 2 illustrates schematically how the drain voltage is apportioned. The
suggested value of V
R
is 130 V: it leads to a maximum drain voltage slightly exceeding 500 V in 220V
AC
or WRM applications, and about 320 V in 110 V
AC
application, thus leaving enough room for an efficient
leakage inductance demagnetization (see below). The maximum duty cycle will be about 60% in
110V
AC
and WRM applications, and close to 36% in 220 V
AC
applications.
Figure 2. Drain voltage composition.
Clamp Diode
forward recovery
700 V
≤
650 V
504 V
317 V
Leak. Inductance resonates
with drain capacitance
Leak. Inductance
demagnetization
margin
V
spike
Transformer
demagnetised
Current flows at the
secondary side
V
R
374 V
187 V
Prim. Inductance resonates
with drain capacitance
Vin
ON
OFF
s
Leakage inductance overvoltage.
The energy stored in the mutual inductance of the transformer at the
primary side is not completely transferred to the secondary, after MOSFET turn-off, until the leakage
inductance is demagnetized. This delays and makes inefficient the energy transfer from primary to sec-
ondary. To minimize this noxious effect the voltage across the leakage inductance (the leakage induct-
ance spike) that resets the inductance itself should be as high as possible. Obviously, this is limited by
the maximum allowable drain voltage. With the reflected voltage selected as previously discussed, it is
possible to allow about 140 V extra voltage in 220 V
AC
or WRM applications and much more in 110 V
AC
applications (see fig. 2). This will affect the design of the clamp circuit.
Transformer efficiency.
By definition, it is the ratio of the power delivered by the secondary winding to
the power entering the primary. The secondary power includes the converter output power and the one
dissipated in the secondary rectifier. Besides the secondary one, the primary power includes the one
dissipated inside the transformer and that not transferred to the secondary side and dissipated on the
leakage inductance. For typical transformers used in converters based on the L6590 family IC's, typical
values of efficiency ranges between 88% and 95%, depending on the power level and on the construc-
tion technique. Efficiency increases with the power level and by using winding interleaving construction
technique. For consistency, check that the input power of the transformer be less than the converter
input power.
Device supply voltage.
The supply voltage range of the IC spans from 7 to 16.5 V. Such a wide range
is envisaged to accommodate the variation that the voltage generated by the self-supply winding may
s
s
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AN1262 APPLICATION NOTE
experience in converters with opto-isolated feedback. This variation is a result of the poor magnetic cou-
pling with the secondary winding. It is then recommended to design the turns ratio of the self-supply
winding so as to get a voltage approximately in the middle of this range (e.g. 11-12 V). This will give
allowance for increasing at heavy load and dropping at zero load.
s
Secondary diode forward drop.
The type of secondary diode will be selected basically depending on the
output voltage. In fact this determines the maximum reverse voltage applied to the diode while the MOS-
FET is switched on. For low output voltages
≤15
V) a Schottky diode can be used and a typical forward
drop of 0.5V can be considered; for higher output voltages an ultrafast PN diode will be used, with a
typical forward drop of 0.8 V.
Bridge Rectifier + EMI filter voltage drop.
This drop is subtracted to the peak of the input AC voltage and
affects the peak voltage of the ripple superimposed on top of the DC voltage across the input bulk ca-
pacitor. A typical value can be 3 V.
PRELIMINARY CALCULATIONS (STEP 1)
s
4
There are a few quantities that need to be calculated before starting the individual design of each functional
block of the converter. They are summarized in table 3.
Table 3. Preliminary calculations (step 1).
Symbol
P
in
Parameter
Converter Input Power
Definition
P
o utmax
P
in
= --------------------
-
η
P
outmax
I
o ut
= --------------------
-
V
ou t
V
PKmin
= V
ACmin
·
2
–
V
BF
I
out
DC Output Current
V
PKmin
V
PKmax
Minimum Peak Input Voltage
Maximum Peak Input Voltage
V
PKmax
=
V
A Cma x
⋅
2
5
BRIDGE RECTIFIER SELECTION
Due to the limited power range that the device is able to handle, no special considerations are needed to select the
diodes of the bridge rectifier. Any 1A rated standard diodes with 400/600 V reverse voltage are suitable. Some man-
ufacturers make integrated bridge rectifiers housed in small packages. See table 4 for some suggested parts.
Table 4. 1A standard silicon rectifier and bridge selection
Type
Diode
Diode
Bridge
Bridge
Bridge
Bridge
Bridge
Bridge
Part Number
1N4004
1N4005
DF04M
DF06M
KBP104G
KBP105G
DFS04M
DFS06M
Rated Voltage
400
600
400
600
400
600
400
600
Package
DO41
DO41
DIL4
DIL4
SIL4
SIL4
DIL4 (SMD)
DIL4 (SMD)
Manufacturer(s)
GI, GS, FAGOR, HTA, ON, TSC
GI, GS, FAGOR, HTA, ON, TSC
GI,TSC
GI,TSC
TSC
TSC
HTA
HTA
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